Method and device for active impedance matching

ABSTRACT

Dynamic matching of a source impedance to a load impedance or the complex conjugate of the load impedance. An embodiment of the invention is a device for active impedance matching comprising a voltage driver having an output connected to a load, means for detecting an output current from the voltage driver to the load, means for scaling the detected output current by a scaling value, and means for subtracting a value representing the scaled detected output current from an input signal of the voltage driver. Another embodiment is a device for active impedance matching comprising a current driver having an output connected to a load, means for detecting an output voltage from the current driver to the load, means for scaling the detected output voltage by a scaling value, and means for subtracting a value representing the scaled detected output voltage from an input signal of the current driver.

This application claims the benefit of U.S. Provisional Application No.60/185,656, filed on Feb. 29, 2000.

BACKGROUND

The invention relates generally to transmission lines and, moreparticularly, to adjusting the terminating and driving impedance of atransmission line to match the characteristic impedance of thetransmission line.

It is well known to skilled practitioners in the electrical arts that ifa source impedance is matched to a complex conjugate of a loadimpedance, maximum power transfer between the source and the load isachieved. However, it is difficult to match the imaginary part of thecomplex impedance and half of the power is lost in the matched sourceimpedance when using passive components for impedance matching. Althoughthis is a characteristic of many electrical circuits, it may take ongreater significance where transmission lines are considered. Withtransmission lines, the primary objective is to avoid reflections in thetransmission line, so the characteristic impedance is assumed to beresistive.

Transmission lines, where the transmission line length is large withrespect to the wavelength of the lowest transmission frequency, arecommonly used for transmission of data between two or more locations. Itis well known in the art of transmission lines, and particularlytransmission lines for transmitting information at high data rates, thatin order to maximize the efficiency of information transfer with minimumloss and dispersion effects, the terminating impedance of a receiver andthe driving impedance of a transmitter must match the characteristicimpedance Z₀ of the transmission line over the frequency range ofinterest. That is, it is desirable to maintain a uniform characteristicimpedance Z₀ along the length of the signal carrying line. Any mismatchin the characteristic impedance across interconnect interfaces willcause reflection of the signal at the interface, resulting in losses anddistortion of the signal in the form of attenuation, echo andcross-talk. Furthermore, multiple reflections from multiple interfacesonly compound the deleterious affect on the information-carrying signal.The classical solution to the impedance matching problem involvesattempting to match the distributed-parameter impedance of thetransmission line with lumped-parameter impedances of resistor,capacitor and inductor circuit elements.

Wide band communication channels, like ADSL modulation over telephoneconventional lines or other wideband modulation schemes, requirematching of line impedances that are complex, where amplitude and phaseare dependent on frequency. Telephone subscriber loops with bridged tapspresent impedance variations at the receiver end that are difficult tomatch using simple circuits. Furthermore, the impedances variations maychange from loop to loop, making it impossible to design a matchingcircuit using generic discrete circuit components. The use offull-duplex techniques, where bi-directional transmission is conductedconcurrently only further complicates the difficulty of matchinginterface impedances to the characteristic impedance of the transmissionline.

There have been a number of different approaches to solving thecharacteristic impedance matching problem. In the most simple andrudimentary form, fixed resistor elements are connected across thetransmission line interfaces to match the interface impedance with thecharacteristic impedance of the transmission line. More compleximpedance matching circuits using combinations of resistor and capacitorelements are often found connected to transmission lines. Impedancematching circuits using passive components may dissipate half of theavailable power at the transmitter, oftentimes reducing its dynamicrange by half. Although power is seldom a major consideration on astandard data transmission line, loss in dynamic range can result inexcessive signal clipping with high peak to average ratios that aretypical of Quadrature Amplitude Modulated signals and DiscreteMulti-Tone signals, used in many modern data transmission systems.

One of the oldest and widely used approaches to match atransmitter-receiver to a transmission line is a hybrid circuit thatmakes use of two transformers and a balance impedance network Z_(L)that, when matched to the characteristic impedance Z₀ of thetransmission line, results in very high isolation between transmitterand receiver circuits. This circuit provides a line termination thatmatches the characteristic impedance of the line and results in noreduction in dynamic range. However, only half the power delivered bythe transmitter is sent to the transmission line, the other half beingwasted on the balancing impedance network Z_(L). In addition to loss oftransmitted power, the balancing impedance network Z_(L) cannotperfectly match a line with bridged taps or multiple interfaces. It isimpractical to add switching circuits to adapt the impedance todifferent lines, where each line has a different configuration of tapsor interfaces along the length of the line. Furthermore, this hybridcircuit makes use of multiple magnetic circuits that have inherentnon-linear characteristics that produce distortion, which adverselyaffects signals with high peak to average ratios. These transformersalso exhibit parasitic capacitance and leakage inductance that mayimpair circuit operation and reduce useful bandwidth.

Another approach that has received increased interest is the use of adifferential driver circuit having two outputs, where each output isconnected through an impedance matching resistor to each of the twoterminals, respectively, of the primary winding of a transformer. Thesecondary winding of the transformer is connected to the transmissionline. However, not only is half of the transmitter power dissipated inthe two impedance matching resistors, but half of the signal amplitudeis also dropped across these resistors. This results in reducing thedynamic range of the signal at the transmitter by one-half and reducingthe maximum power available to drive the transmission line byone-fourth. The transformer provides for scaling the line impedance tocompensate for this reduction and for generating enough peak voltagewithout excessive clipping. Two amplifiers, each connected across aterminating resistor receive the signal on the transmission line. Thiscircuit may only perform better than the hybrid circuit described abovein the high frequency range, where the line impedance will be mostlyresistive in nature. Although more complex networks may replace theseterminating resistors, the resultant configuration would also sufferfrom the same limitations as the hybrid circuit described above, namelylow power efficiency and reduced dynamic range.

All of these solutions assume that the characteristic impedance of thetransmission line is fixed and known, and therefore terminatedaccordingly. These solutions result in reduced power available to thetransmission line, reduced dynamic range of the signal, and losses anddistortion in the signal. Although more pronounced with transmissionlines, these problems apply to many electrical circuits.

For the foregoing reasons, it is desirable to have a method and devicefor driving and receiving signals on a transmission line that does notexhibit loss of the available transmitter power to drive the line, doesnot suffer from a reduction in dynamic signal range, and dynamicallymatches the driving and terminating impedance at the interfaces to thecharacteristic impedance of the transmission line.

SUMMARY

The present invention is directed to a method and device for driving aload with active impedance matching that satisfies these needs. Thepresent invention is particularly suitable for providing a method anddevice for driving and receiving signals on a transmission line thatdoes not exhibit loss of the available transmitter power to drive theline, does not suffer from a reduction in dynamic signal range, anddynamically matches the transmission line interface driving andterminating impedance to the characteristic impedance of thetransmission line.

In a voltage driver version of the present invention, a means isprovided for sensing the current provided to a load by a voltage source,and the magnitude of the voltage source is automatically adjusted bynegatively feeding back a voltage to an input that represents a scaledvalue of the sensed current multiplied by an impedance that matches theload impedance. The result is a voltage source having an effectiveinternal impedance that matches the load impedance, but yet maintainsfull dynamic signal range without a loss of transmitted power to theload.

In a current driver version of the present invention, a means isprovided for sensing the voltage provided to a load by a current source,and the magnitude of the current source is automatically adjusted bynegatively feeding back a current to an input that represents a scaledvalue of the sensed voltage divided by an impedance that matches theload impedance. The result is a current source having an effectiveinternal impedance that matches the load impedance, but yet maintainsfull dynamic signal range without a loss of transmitted power to theload.

Although the present method and device is applicable to many electricalcircuits, its application is particularly suitable to transmissionlines.

A device having features of the present invention is a device withactive impedance matching for driving a load that comprises a voltagedriver having an output connected to a load, means for detecting anoutput current from the voltage driver to the load, means for scalingthe detected output current by a scaling value, and means forsubtracting a value representing the scaled detected output current froman input signal of the voltage driver. The means for scaling thedetected output current may be a multiplier having an input comprisingthe detected output current and another input comprising the scalingvalue, an output of the multiplier representing the scaled outputcurrent. The device of claim 2, wherein the scaling value is a valuerepresenting a load impedance to be matched. The means for scaling thedetected output current may be an amplifier having an input comprisingthe detected output current and a gain equal to the scaling value, anoutput of the amplifier representing the scaled output current. Themeans for detecting an output current may be a transformer having aprimary winding in series with the output current. The means fordetecting an output current may be a resistor in series with the outputcurrent and an amplifier with inputs connected to terminals of theresistor. The means for subtracting may be a summing junction of anoperational amplifier. The load may be a transmission line. The scalingvalue may be a characteristic impedance of the transmission line. Themeans for scaling and the means for subtracting may comprise a digitalsignal processor.

In an alternative embodiment of the present invention, a device withactive impedance matching for driving a load comprises a current driverhaving an output connected to a load, means for detecting an outputvoltage from the current driver to the load, means for scaling thedetected output voltage by a scaling value, and means for subtracting avalue representing the scaled detected output voltage from an inputsignal of the current driver. The means for scaling the detected outputvoltage may be a multiplier having an input comprising the detectedoutput voltage and another input comprising the scaling value, an outputof the multiplier representing the scaled output voltage. The scalingvalue may be a value representing a load impedance to be matched. Themeans for scaling the detected output voltage may be an amplifier havingan input comprising the detected output voltage and a gain equal to thescaling value, an output of the amplifier representing the scaled outputvoltage. The means for detecting an output voltage may be an amplifierwith inputs connected to the outputs of the current driver. The meansfor detecting an output voltage may be a transformer with primaryterminals connected to the outputs of the current driver. The means forsubtracting may be a summing junction of an operational amplifier. Theload may be a transmission line. The scaling value may be acharacteristic impedance of the transmission line. The means for scalingand the means for subtracting may comprise a digital signal processor.

In another alternative embodiment of the present invention, a method fordriving a load with active impedance matching, comprises connecting anoutput of a voltage driver to a load, detecting an output current fromthe voltage driver to the load, scaling the detected output current by ascaling value, and subtracting a value representing the scaled detectedoutput current from an input signal of the voltage driver. Scaling thedetected output current may comprise multiplying the detected outputcurrent by the scaling value, an output of the multiplicationrepresenting the scaled output current. The scaling value may be a valuerepresenting a load impedance to be matched. The detected output currentmay comprise amplifying the detected output current by the scaling valuefor obtaining a value representing the scaled output current. Detectingan output current may comprise connecting a primary winding of atransformer in series with the output current. Detecting an outputcurrent may comprise connecting a resistor in series with the outputcurrent and connecting inputs of an amplifier to terminals of theresistor. Subtracting may comprise summing currents into a summingjunction of an operational amplifier. The load may be a transmissionline. The scaling value may be a characteristic impedance of thetransmission line. Scaling and subtracting may comprise processinginstructions of a digital signal processor.

In another alternative embodiment of the present invention, a method fordriving a load with active impedance matching comprises connecting anoutput of a current driver to a load, detecting an output voltage fromthe current driver to the load, scaling the detected output voltage by ascaling value, and subtracting a value representing the scaled detectedoutput voltage from an input signal of the current driver. Scaling thedetected output voltage may comprise multiplying the detected outputvoltage by the scaling value, an output of the multiplicationrepresenting the scaled output voltage. The scaling value may be a valuerepresenting a load impedance to be matched. Scaling the detected outputvoltage may comprise amplifying the detected output voltage by thescaling value for obtaining a value representing the scaled outputvoltage. Detecting an output voltage may comprise connecting inputs ofan amplifier to outputs of the current driver. Detecting an outputvoltage may comprise connecting a primary winding of a transformer tothe outputs of the current driver. Subtracting may comprise summingcurrents into a summing junction of an operational amplifier. The loadmay be a transmission line. The scaling value may be a characteristicimpedance of the transmission line. Scaling and subtracting may compriseprocessing instructions of a digital signal processor.

In another alternative embodiment of the present invention, a method fordriving a load with active impedance matching comprises connecting anoutput of a voltage driver to a load, detecting an output current valuefrom the voltage driver to the load, connecting the detected outputcurrent to an analog-to-digital converter, converting the detectedoutput current value to a digital representation by theanalog-to-digital converter, connecting the digital representation ofthe output current at an output of the analog-to-digital converter to aninput of a digital signal processor, connecting a digital representationof an input signal to another input of the digital signal processor,executing a program in the digital signal processor, providing andigital representation output from the digital signal processor to adigital-to-analog converter, and connecting an output of thedigital-to-analog converter to an input of the voltage driver. Themethod may further comprise interposing an anti-aliasing low-pass filterbetween the detected current output and the analog-to-digital converter.The method may further comprise interposing an interpolation low-passfilter between the output of the digital-to-analog converter and theinput of the voltage driver. The step of connecting a digitalrepresentation of an input signal may comprise connecting an inputsignal to another input of the voltage driver. The step of executing aprogram in the digital signal processor may further comprise executingan initialization routine, reading an input voltage value, associating atime value with the input voltage value, adjusting the time value with atime domain filter delay, reading an output current value from theanalog-to-digital converter, applying the output current value to thetime domain filter, subtracting the filtered output current value fromthe adjusted input voltage value, outputting the result of thesubtraction to a digital-to-analog converter, repeating steps b. throughh. if the program is not terminated, and ending the process if theprogram is terminated.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects, and advantages of the presentinvention will become understood with regard to the followingdescription, appended claims, and accompanying drawings where:

FIG. 1A shows a half duplex configuration of a transmitter/receivercircuit;

FIG. 1B shows a full duplex configuration of a transmitter/receivercircuit;

FIG. 2A shows a Thevenin equivalent of a voltage transmitter circuit;

FIG. 2B shows an equivalent circuit of the circuit shown in FIG. 2A;

FIG. 3A shows a Norton equivalent of a current transmitter circuit;

FIG. 3B shows an equivalent circuit of the circuit shown in FIG. 3A;

FIG. 4 shows a block diagram of a voltage transmitter circuit;

FIG. 5 shows a block diagram of a current transmitter circuit;

FIG. 6A shows a circuit diagram of an a voltage transmitter using aninductive sensor;

FIG. 6B shows a circuit diagram of a voltage transmitter using aresistive sensor;

FIG. 7 shows a circuit diagram of a DSP implementation of a voltagetransmitter;

FIG. 8A shows a flow diagram of a program executed in the DSP of FIG. 7;and

FIG. 8B shows a flow diagram of an initialization routine.

DETAILED DESCRIPTION

Turning now to FIG. 1A, FIG. 1A shows a half duplex configuration 10 oftransmitter/receiver circuit connected to a transmission line 140. Thisconfiguration is well known to skilled practitioners in the relevantart. In the half duplex configuration of FIG. 1A, a transmit inputsignal 112 is connected to an input of a differential transmitter 102.An output of the transmitter is connected to a switch 106. A receiveoutput signal 114 is provided by an output of a differential receiver104. An input of the receiver 104 is connected to the switch 106. Atransmission line signal 108, 110 is connected to the switch 106 suchthat the when the switch 106 is in position A, the transmission linesignal 108, 110 is connected to the output of the transmitter 102, fortransmitting a transmission line signal 108, 110. Alternatively, whenthe switch is in position B, the transmission line signal 108, 110 isconnected to the input of the receiver 104, for receiving a transmissionline signal 108, 110. Although the switch 106 is depicted as anelectromechanical device, skilled practitioners would recognize that asemiconductor device would normally provide this function for halfduplex operation. In half duplex operation, a transmission line 140 istransmitting signals in one direction at a time.

Turning now to FIG. 1B, FIG. 1B shows a full duplex configuration 12 oftransmitter/receiver circuit connected to a transmission line 140. Thisconfiguration is also well known to skilled practitioners in therelevant art. In the half duplex configuration of FIG. 1B, a transmitinput signal 112 is connected to an input of a differential transmitter102. An output of the transmitter is connected to a transmission linesignal 108, 110 through one port 128 of a hybrid coil or four-to-twowire converter 126. A receive output signal 114 is provided by an outputof a differential receiver 104. An input of the receiver 104 isconnected the transmission line signal 108, 110 through another port 130of the hybrid coil or four-to-two wire converter 126. In this fullduplex configuration 12, the transmission line 140 may transmit signalsin both directions simultaneously, the directional coupling and linetermination being performed by the hybrid coil or four-to-two wireconverter 126.

The subsequent descriptions of embodiments of the present inventionpertain to the transmitter 102 of FIG. 1A and FIG. 1B. Skilledpractitioners will recognize that embodiments of the present inventionmay be used with a receiver 104 in either a half duplex configuration ofFIG. 1A or the full duplex configuration of FIG. 1B, eliminating theneed for the switch 106 or the hybrid coil or four-to-two wireconverter.

Turning now to FIG. 2A and FIG. 2B, FIG. 2A shows a Thevenin equivalentcircuit 20 of a voltage transmitter circuit and FIG. 2B shows anequivalent circuit 22 of the circuit 20 shown in FIG. 2A. In FIG. 2A, avoltage generator V_(g) 202 represents a Thevenin equivalent opencircuit voltage source and an impedance Z_(g) 204 represents a Theveninequivalent impedance. The circuit has an output voltage V_(o) 208 and anoutput current I_(o) 206 connected to a load impedance Z_(L) 240. Bymeasuring the output current I_(o) 206 and negatively feeding it backwith an appropriate gain required to synthesize the impedance Z_(g) 204,the equivalent circuit 22 shown in FIG. 2B is formed. FIG. 2B comprisesa voltage generator 222 having a value of

V _(o) =V _(g) −Z _(g) I _(o)

that provides the output voltage V_(o) 208 and the output current I_(o)206 connected to a load impedance Z_(L) 240. Note that V_(o) 208 andI_(o) 206 are the same in FIG. 2A and FIG. 2B. If the value of V_(g) isset to zero (short-circuit) and a current generator of unity value isconnected to the outputs of the circuits shown in both FIG. 2A and FIG.2B, the value of the voltage V_(o)=Z_(g) is the sane in both circuits.This example illustrates the principle of operation of one of theembodiments of the present invention. That is, in a voltage transmittercircuit, the source impedance Z_(g) 204 may be matched to a loadimpedance Z_(L) 240 by measuring the output current from the circuit andnegatively feeding back a scaled part of the output current determinedby the value of the load impedance Z_(L) 240. In this manner, maximumpower transfer may be achieved by setting Z_(g) 204=Z_(L) 240 withoutpower loss in Z_(g) 204.

Turning now to FIG. 3A and FIG. 3B, FIG. 3A shows a Norton equivalentcircuit 30 of a current transmitter circuit and FIG. 3B shows anequivalent circuit 32 of the circuit 30 shown in FIG. 3A. In FIG. 3A, acurrent generator I_(g) 302 represents a Norton equivalent short circuitcurrent source and an impedance Z_(g) 304 represents a Norton equivalentimpedance. The circuit has an output voltage V_(o) 308 and an outputcurrent I_(o) 306 connected to a load impedance Z_(L) 340. By measuringthe output voltage V_(o) 306 and negatively feeding it back with anappropriate gain required to synthesize the admittance 1/Z_(g) 304, theequivalent circuit 32 shown in FIG. 3B is formed. FIG. 3B comprises acurrent generator 322 having a value of

I _(o) =I _(g) −V _(o) /Z _(g)

that provides the output voltage V_(o) 308 and the output current I_(o)306 connected to the load impedance Z_(L) 340. Note that V_(o) 308 andI_(o) 306 are the same in FIG. 3A and FIG. 3B. If the value of I_(g) isset to zero (open-circuit) and a voltage generator of unity value isconnected to the outputs of the circuits shown in both FIG. 3A and FIG.3B, the value of the current I_(o)=1/Z_(g) is the same in both circuits.This example illustrates the principle of operation of one of theembodiments of the present invention. That is, in a current transmittercircuit, the source impedances Z_(g) 304 may be matched to a loadimpedance Z_(L) 340 by measuring the output voltage from the circuit andnegatively feeding back a scaled part of the output voltage, determinedby the value of the load admittances 1/Z_(L) 340. In this manner,maximum power transfer may be achieved by setting Z_(g) 304=Z_(L) 340without power loss in Z_(g) 304.

Turning now to FIG. 4, FIG. 4 shows a block diagram of a voltagetransmitter circuit 40 connected to a transmission line 440 having acharacteristic impedance Z_(o) 442. The block diagram 40 illustrates ause of current feedback 418 from the an output of a voltage driver 410to synthesize a driver circuit whose Thevenin equivalent is a voltagegenerator of amplitude 2V_(in) in series with an impedance Z_(o),similar to the circuit shown in FIG. 2A. The input voltage V_(in) 402 issummed with a negative feedback voltage V_(fb) 420 to provide an inputvoltage of V_(in)=V_(fb) to the voltage driver 410. Since the voltagedriver 410 has a voltage gain of two, the output voltage V_(o) 408 ofthe voltage driver 410 is V_(o)=2V_(in)−2V_(fb). A transformer 412having a turns ratio of n senses the output current I_(o) 406 andprovides the signal I_(o)/n to one input to a multiplier 414. Anotherinput signal to the multiplier is the constant value nZ_(o)/2 516.Therefore, the output signal of the multiplier 414 isV_(fb)=(I_(o)/n)(nZ_(o)/2)=I_(o)Z_(o)/2 420. By substituting this valueof V_(fb) 420 into the expression above for the output voltage V_(o)408, the output voltage

V _(o)=2V _(in) −I _(o) Z _(o)

This expression for the output voltage V_(o) 408 has the form of theoutput voltage of FIG. 2A and FIG. 2B, and illustrates how the drivingsource impedance may be matched to the characteristic impedance of atransmission without the use of power consuming components. Summarizing,the output current 406 is measured, scaled and multiplied by animpedance nZ_(o)/2, resulting in the feedback voltage V_(fb) 420. Thefeedback voltage V_(fb) 420 is subtracted from the input voltage V_(in)402 and fed to an input of the voltage driver 410, which has a gain oftwo. The scaling and multiplication may be accomplished on acurrent-to-voltage converter, the output driver, or through use ofdigital filtering techniques in a Digital Signal Processor (DSP). Notethat the voltage driver 410 will only generate the voltage seen by theline V_(o), even if the Thevenin equivalent circuit has a voltagegenerator of twice this value. This method achieves the objective ofimpedance matching without wasting power or dropping a voltage in animpedance-matching resistor. This method is also adaptable to the use ofa DSP to enable more accurate and adaptive matching through digitalsignal processing techniques. It also allows full duplex communicationover the same transmission line.

Turning now to FIG. 5, FIG. 5 shows a block diagram of a currenttransmitter circuit 50 connected to a transmission line 540 having acharacteristic impedance Z_(o) 542. The block diagram 50 illustrates ause of output voltage feedback V_(o) 508 from an output of atransconductance driver 510 to synthesize a driver circuit 50 whoseNorton equivalent circuit is a current generator of amplitudeI_(o)=kV_(in) 506 having an internal shunt impedance Z_(g)=Z_(o), thecharacteristic impedance of the line, and a transconductance of k,similar to FIG. 3A. The input voltage V_(in) 502 is summed with anegative feedback voltage V_(fb) 520 to provide an input voltage ofV_(in)−V_(fb) to the transconductance driver 510. Since thetransconductance driver 510 has a transconductance of 2k, the outputcurrent I_(o) 506 of the transconductance driver 510 isI_(o)=2kV_(in)−2kV_(fb). An amplifier 512 senses the output voltageV_(o) 508 and provides this signal V_(o) 508 to one input of amultiplier 514. Another input signal to the multiplier is the constantvalue 1/2kZ_(o) 516. Therefore, the output signal of the multiplier 514is V_(fb)=V_(o)/2kZ_(o) 520. By substituting this value of V_(fb) 520into the expression above for the output current I_(o) 506, the outputcurrent

I _(o)=2kV _(in) −V _(o) /Z _(o)

This expression for the output current I_(o) 506 has the form of theoutput current of FIG. 3A and FIG. 3B, and illustrates how the drivingsource impedance may be matched to the characteristic impedance of atransmission without the use of power consuming components. Summarizing,the output voltage 508 is measured and multiplied by an admittance1/2kZ_(o), resulting in the feedback voltage V_(fb) 520. The feedbackvoltage V_(fb) 520 is subtracted from the input voltage V_(in) 502 andfed to an input of the transconductance driver 510, which has a gain of2k, where k is the transconductance of the transconductance driver 510.The scaling and multiplication may be accomplished on an amplifier, theoutput driver, or through use of digital filtering techniques in aDigital Signal Processor (DSP). This method achieves the objective ofimpedance matching without wasting power or dropping a voltage in animpedance-matching resistor. This method is also adaptable to the use ofa DSP to enable more accurate and adaptive matching through digitalsignal processing techniques. It also allows full duplex communicationover the same transmission line.

Turning now to FIG. 6A, FIG. 6A shows a circuit diagram 60 of a voltagetransmitter circuit using an inductive sensor connected to atransmission line 640 having a characteristic impedance Z_(o) 642. Thevoltage transmitter circuit of FIG. 6A comprises a first operationalamplifier 614 and a second operational amplifier 616 having input andoutputs connected to a resistor network 618. An input voltage V_(in) 602is connected to one terminal of an input impedance nZ_(o)/4 603. Anotherterminal of the input impedance nZ_(o)/4 603 connects a summing junction620. An output voltage V_(o) 608 is derived between an output terminalof the first amplifier 614 and an output terminal of the secondamplifier 616. The voltages at the output terminals of the amplifiersare mirror images of each other. That is, when the output terminal ofthe first amplifier is at given voltage, the output terminal of thesecond amplifier is at an equal voltage of opposite polarity. Themagnitude of the voltage at the output terminal of each amplifier 614,616 is V_(o)/2. A terminal of a feedback impedance nZ_(o)/2 604 isconnected to the output terminal of the second amplifier 616 and anotherterminal of the feedback impedance nZ_(o)/2 604 is connected to thesumming junction 620. A transformer 612 senses the output current I_(o)606 and provides a scaled feedback current I_(fb)=I_(o)/n 622 to thesumming junction 620. By summing the currents into the summing junction,an expression for the output voltage may be derived

V _(o)=4V _(in) −I _(o) Z _(o)

This expression for the output voltage V_(o) 608 has the form of theoutput voltage of FIG. 2A and FIG. 2B, and illustrates how the drivingsource impedance may be matched to the characteristic impedance of atransmission without the use of power consuming components.

Turning now to FIG. 6B, FIG. 6B shows a circuit diagram 65 of a voltagetransmitter circuit using a resistive sensor connected to a transmissionline 690 having a characteristic impedance Z_(o) 692. FIG. 6B is similarto FIG. 6A, except that a current sensing resistor R_(s) 662, adifferential amplifier 660 with a gain of A and a resistor nR_(s)/A 674have replaced the current sensing transformer of FIG. 6A. The inputs ofthe amplifier 660 are connected to the terminals of the sensing resistorR_(s) 662. The output current I_(o) 656 through the sensing resistorR_(s) 662 creates a voltage that is detected by the amplifier 660. Theoutput of the amplifier 660 is connected to a terminal of the resistornR_(s)/A 674 and another terminal of the resistor nR_(s)/A 674 isconnected to a summing junction 670. The voltage transmitter circuit ofFIG. 6B further comprises a first operational amplifier 664 and a secondoperational amplifier 666 having input and outputs connected to aresistor network 668. An input voltage V_(in) 652 is connected to oneterminal of an input impedance nZ_(o)/4 653. Another terminal of theinput impedance nZ_(o)/4 653 connects the summing junction 670. Anoutput voltage V_(o) 658 is derived between an output terminal of thefirst amplifier 664 and an output terminal of the second amplifier 666.The voltages at the output terminals of the amplifiers are mirror imagesof each other. That is, when the output terminal of the first amplifieris at given voltage, the output terminal of the second amplifier is atan equal voltage of opposite polarity. The magnitude of the voltage atthe output terminal of each amplifier 664, 666 is V_(o)/2. A terminal ofa feedback impedance nZ_(o)/2 654 is connected to the output terminal ofthe second amplifier 666 and another terminal of the feedback impedancenZ_(o)/2 654 is connected to the summing junction 670. By summing thecurrents into the summing junction, an expression for the output voltagemay be derived:

 V _(o)=4V _(in) −I _(o) Z _(o)

This expression for the output voltage V_(o) 658 has the form of theoutput voltage of FIG. 2A and FIG. 2B, and illustrates how the drivingsource impedance may be matched to the characteristic impedance of atransmission without the use of power consuming components.

Turning now to FIG. 7, FIG. 7 shows a circuit diagram 70 of a DSPimplementation of a voltage transmitter connected to a transmission line740 having a characteristic impedance Z_(o) 742. The amplifiers 714,716, resistor network 718, input resistor 703, feedback resistor 704,and transformer 712 are similar to those corresponding elements shown inFIG. 6A. The voltage transmitter circuit of FIG. 7 comprises a firstoperational amplifier 714 and a second operational amplifier 716 havinginput and outputs connected to a resistor network 718. An input resistor703 connects between an output of an interpolation filter 728 and asumming junction 734. A feedback resistor 704 connects between thesumming junction 734 and an output of the second amplifier. An outputvoltage V_(o) 708 is derived between an output terminal of the firstamplifier 714 and the output terminal of the second amplifier 716. Thevoltages at the output terminals of the amplifiers are mirror images ofeach other. That is, when the output terminal of the first amplifier isat given voltage, the output terminal of the second amplifier is at anequal voltage of opposite polarity. The magnitude of the voltage at theoutput terminal of each amplifier 714, 716 is V_(o)/2. A transformer 712senses the output current I_(o) 706 and provides a scaled feedbackcurrent I_(o)/n to the summing junction 736 of an I/V converter 730. Afeedback resistor 732 connects between an output of the I/V converter730 and the summing junction 736 of the I/V converter 730. The output ofthe I/V converter 730 is connected to the input of an anti-aliasingfilter 720. An output of the anti-aliasing filter 720 is connected to aninput of an analog-to-digital (A/D) converter 722. Outputs from the A/Dconverter 722 are connected to a DSP 724. Outputs from the DSP 724 areconnected to the inputs of a digital-to-analog (D/A) converter 726. Anoutput from the D/A converter 726 is connected to the interpolationfilter 728. Normally the DSP generates the signals to be transmittedover the transmission line, functioning as a modem. Alternatively, adigital input voltage V_(in) 702 is connected an input terminal of theDSP. By performing scaling and feedback functions in a DSP 724,intelligence is added to the process that allows sophisticated andadaptive matching of the characteristic impedance Z_(o) of thetransmission line. The DSP 724 may send a voltage signal V_(o) to theline, measure the resulting current and calculate a transfer function,such as nV/I. With sufficient over-sampling to avoid excessive phaseshift, the line impedance may be matched by multiplying the line currentI_(o) by a suitable transfer function and subtracting the result fromtwice the intended output signal V_(o). For full duplex operation, thereceived signal may be obtained by digitally subtracting the transmittedsignal from the line voltage V_(o) measured by a receiver. Since theinvention requires the use of line impedance models, with a DSP, thesemodels are no longer limited to simple passive network elements.

Turning now to FIG. 8A, FIG. 8A shows a representative flow diagram 80of a program executed in the DSP of FIG. 7. The DSP is started 802whenever it is initially powered on or reset. A first step is anexecution of an initialization routine 804. The details of theinitialization routine 804 are described in the description of FIG. 8B.The DSP then reads a value representing an input voltage 806, associatesa current time value with the input voltage 808, and adjusts the timevalue for a time domain filter delay 810. Concurrently with these steps,the DSP reads a value representing an output current I_(o)/n from an A/Dconverter 812, calculates an error from a predicted current and updatestime domain filter 813, and applies the output current value to anZ_(o)/2 time domain filter 814. The DSP then subtracts the filteredoutput current value from the input voltage value 816, and provides theresultant value to a D/A converter. If the DSP operation is to beterminated 820, the process is ended 822. If not terminated 820, theprocess beginning with concurrently reading input voltage values 806 andreading output current values 812 is repeated. As an alternative to theinitialization routine 804 described in FIG. 8B, the initializationroutine may be limited to setting initial parameters of the time domainfilter for synthesizing an output impedance of approximate value. Then,referring to FIG. 8A, the DSP would read the output current 812,calculate an error from a predicted current and update the time domainfilter 813 with a fraction of the error to improve the matching in arecursive manner. These updated values would then be used to adjust theoutput voltage 816.

Turning now to FIG. 8B, FIG. 8B shows a flow diagram 85 of aninitialization routine depicted as step 804 in FIG. 8A. If the DSPrequires initialization, as described in the description of FIG. 8A, theinitialization routine is started 850. A value of the characteristicimpedance Z_(o) of the transmission line is set to approximately matchthe transmission line and this value is applied to a time domain filter852. For example, an approximate value of 600 ohms is used for telephonelines, 120 ohms for twisted pair, or 50 ohm for coaxial cable. The DSPthen initiates a request to a receiver at the opposite end of thetransmission line to present a short-circuit for a fixed amount of time854, simulates a short-circuit output by setting an output voltage to aconstant 856, and measures the value of an output current to find valuesfor a short-circuit impedance Z_(is) versus frequency 858. The DSP theninitiates a request to the receiver at the opposite end of thetransmission line to present an open-circuit for a fixed amount of time860, simulates an open-circuit output by setting an output voltage tozero and setting Z_(o) to a high value 862, and measures the value ofthe output current to find values for an open circuit impedance Z_(io)versus frequency 864. The DSP then computes values for thecharacteristic impedance Z_(o)=(Z_(is)Z_(os))^(½) versus frequency 866and sets time domain filter parameters to match Z_(o) versus frequency869. Control is then returned to the main program 870.

Although the present invention has been described in detail withreference to certain preferred embodiments, it should be apparent thatmodifications and adaptations to those embodiments may occur to personsskilled in the art without departing from the spirit and scope of thepresent invention as set forth in the following claims.

What is claimed is:
 1. A device for active impedance matching,comprising: a. a voltage driver having an output connected to a load; b.means for detecting an output current from the voltage driver to theload; c. means for scaling the detected output current by a scalingvalue; and d. means for subtracting a value representing the scaleddetected output current from an input signal of the voltage driver. 2.The device of claim 1, wherein the means for scaling the detected outputcurrent is a multiplier having an input comprising the detected outputcurrent and another input comprising the scaling value, an output of themultiplier representing the scaled output current.
 3. The device ofclaim 2, wherein the scaling value is a value representing a loadimpedance to be matched.
 4. The device of claim 1, wherein the means fordetecting an output current is a transformer having a primary winding inseries with the output current.
 5. The device of claim 1, wherein themeans for subtracting is a summing junction of an operational amplifier.6. The device of claim 1, wherein the load is a transmission line. 7.The device of claim 6, wherein the scaling value is a characteristicimpedance of the transmission line.
 8. A method for active impedancematching, comprising: a. connecting an output of a voltage driver to aload; b. detecting an output current from the voltage driver to theload; c. scaling the detected output current by a scaling value; and d.subtracting a value representing the scaled detected output current froman input signal of the voltage driver.
 9. The method of claim 8, whereinscaling the detected output current comprises multiplying the detectedoutput current by the scaling value, an output of the multiplicationrepresenting the scaled output current.
 10. The method of claim 9,wherein the scaling value is a value representing a load impedance to bematched.
 11. The method of claim 8, wherein scaling the detected outputcurrent comprises amplifying the detected output current by the scalingvalue for obtaining a value representing the scaled output current. 12.The method of claim 8, wherein detecting an output current comprisesconnecting a primary winding of a transformer in series with the outputcurrent.
 13. The method of claim 8, wherein detecting an output currentcomprises connecting a resistor in series with the output current andconnecting inputs of a differential amplifier to terminals of theresistor.
 14. The method of claim 8, wherein subtracting comprisessumming currents into a summing junction of an operational amplifier.15. The method of claim 8, wherein the load is a transmission line. 16.The method of claim 15, wherein the scaling value is a characteristicimpedance of the transmission line.